1. Field of the Invention
The present invention generally relates to integrated circuit output drivers. More specifically, the invention relates to both a circuit and a method for adjusting signal transmission parameters of a signal designated for transmission from a first integrated circuit to a second integrated circuit.
2. Discussion of the Related Art
Integrated circuits (ICs) are electrical circuits which incorporate transistors, resistors, capacitors, and other components onto a single semiconductor xe2x80x9cchipxe2x80x9d in which the components are interconnected to perform a given function. Typical examples of ICs include microprocessors, programmable logic devices (PLDs), electrically erasable programmable memory devices (EEPROMs), random access memory devices (RAMs), operational amplifiers, voltage regulators, and others. Generally, ICs incorporate chip pins, which are configured for enabling electrical interconnection of external electronic components, such as other ICs, high-power amplifiers, discrete external circuit components, and other similar devices. IC electrical interconnection points may be physically and electrically fixed to a printed circuit board via a plurality of solder contact points or pads, which in turn are electrically coupled to a plurality of printed circuit board conductors commonly known as traces. Alternatively, for higher frequency applications using a flip-chip or bump chip, a plurality of solder columns or solder bumps strategically located on the die may be used to provide the physical and electrical interface between the various circuit components on the die and external circuit elements (i.e., other semiconductor dies, ICs, or other such devices). The printed circuit board traces, or IC packages in the case of high-frequency circuit applications, provide a transmission media for input and output signals to and from each IC. In addition, the printed circuit board traces or IC packages may serve to supply any necessary power and electrical ground references to the ICs.
An exemplary configuration is illustrated in FIG. 1A. In this regard, the figure shows a portion of a printed circuit board 20 having a plurality of contact pads 22 arranged to receive a plurality of ICs 10a-10d. Generally, as illustrated in FIG. 1A, a plurality of IC pins 12 are electrically and physically associated via the plurality of printed circuit pads 22. As further illustrated in FIG. 1A, a printed circuit board trace 25 may be provided along the upper or lower surface of the printed circuit board 20 or between two or more printed circuit board layers routed to one or both surfaces using a via in order to communicatively couple one or more IC pins 12 from a first IC 10a to designated circuits and/or circuit components external to the first IC 10a. For example, in FIG. 1A, the right most pin 12 associated with IC 10a is coupled to the second pin from the left associated with IC 10b via the printed circuit board trace 25.
An IC output driver is typically configured for providing signals designated for transmission to the aforementioned external circuits or circuit components. The IC output driver supplies an amplified version of the signals to be communicated to one or more external devices to a chip pin associated with the IC. It will be appreciated that for high-frequency applications it may be desirable to reduce the number of possible impedance transitions that may confront a particular signal. As previously explained, a semiconductor die may be interconnected to an IC package using a plurality of strategically placed solder columns or solder bumps to physically and electrically connect the various circuits on the die to the IC package. Such an arrangement is illustrated in FIG. 1B. In this regard, FIG. 1B illustrates a cross-sectional view representing the assembly of a flip-chip 10axe2x80x2 to an open cavity ball-grid array 24. As illustrated, the flip-chip 10axe2x80x2 may contain one or more (one shown for simplicity of illustration) contact pads 22xe2x80x2 each having its own solder bump 28. Similarly, the ball-grid array 24 may be configured with one or more spatially separated contact pads 22xe2x80x2 each having its own solder bump 28. The flip-chip 10axe2x80x2 may be placed in substantial contact alignment with the open cavity ball-grid array 24. Heat may then be applied such that the one or both of the solder bumps 28 reaches a melting point. Once the heat is removed and the one or more solder bumps 28 cools, the flip-chip 10axe2x80x2 is both physically and electrically interconnected to the ball-grid array 24. It will be appreciated that internal conductors within the flip-chip 10axe2x80x2 die, the contact pads 22xe2x80x2, the solder bumps 28, along with the associated elements and electrical conductors on the ball-grid array 24 form the transmission media for IC to IC signal transfers. As is known, the ball-grid array 24 may provide a plurality of conductors suitably configured to supply each of the one or more interface signals to pre-designated locations on one or more separate and distinct semiconductor dies.
The block diagram of FIG. 2 further illustrates an IC to IC signal transfer. As presented in FIG. 2, a first IC 10a affixed to the printed circuit board 20 may be electrically coupled to a second IC 10b as follows. An output driver 14 configured to amplify a signal 30 may supply the amplified signal 30 via a first IC pin 12a to a first printed circuit pad 22a. The first printed circuit pad 22a may be electrically coupled to the printed circuit board trace 25, which may be further coupled to a second printed circuit pad 22b. As illustrated in FIG. 2, the second printed circuit pad 22b may be coupled to a second IC pin 12b associated with the second IC 10b. More specifically, the second IC pin 12b may be coupled to a designated receiver 16 within the second IC 10b. As further illustrated in the block diagram of FIG. 2, the IC to IC signal transfer is not point to point limited. In this example, a single output driver 14 to receiver 16 transfer is illustrated. It will be appreciated that a bus 15 may be coupled to the printed circuit board trace 25, which may further distribute the amplified signal 30 to various devices throughout the printed circuit board 20. The distribution of the amplified signal 30 from the first IC 10a to the second IC 10b via the IC pins 12, the printed circuit pads 22, and the printed circuit board trace 25 may be modeled using transmission line theory.
The electrical connection described above with regard to the block diagram of FIG. 2, contains parasitic resistance, inductance, and capacitance, which interfere with the transmission of the signal 30 from the output driver 14 to the receiver 16. The parasitic interference increases the load seen by the output driver 14. Transmission line theory teaches that for transmission lines having a finite length terminated in a non-characteristic impedance, time-varying signals transmitted along the transmission line may suffer from reflected signals. Conversely, for time-varying signals transmitted along transmission lines of a finite length terminated in the characteristic impedance of the transmission line, the reflected signals will vanish.
Impedance mismatches between the output driver 14 and the various signal transmission media of the signal transmission path, as well as, between the receiver 16 and the various signal transmission media of the signal transmission path may produce signal reflections at the output driver end and/or the receiver end of the signal transmission path. These signal reflections may propagate along the transmission path and may potentially result in less than desired system performance. A representative signal 30 including such signal reflections, i.e., reflections 32 and 34, is depicted in FIG. 3. Such reflections may cause additional noise and ringing (i.e., excessive transient voltage swings). Under some impedance mismatch conditions, signal reflections will become so severe as to result in incorrect data transmissions between the output driver 14 (FIG. 2) and the receiver 16 (FIG. 2).
From circuit theory it is known that a maximum transfer of power from a given voltage source to a load occurs when the load impedance is the complex conjugate of the source impedance. In transmission line terminology, a line is xe2x80x9cmatchedxe2x80x9d and is most efficient when the load impedance is equal to the characteristic impedance of the transmission line. As a result, it is highly desirable to closely match the output impedance of the output driver 14 to the various components comprising the conductive pathway, hereinafter the transmission line (e.g., the IC pins 12, the printed circuit pads 22, and the printed circuit board trace 25 of FIG. 2), and the input impedance of the receiver 16. Because of the parasitic resistance, inductance, and capacitance present within the transmission line, the output driver 14 is preferably designed to avoid excessive voltage swings when switching occurs (particularly for high speed or low-power I/O signal transmissions). Generally, output drivers 14 are designed for an output impedance of 50 Ohms to match the characteristic impedance of the printed circuit board trace 25 (FIGS. 1 and 2), which will transfer the signal to various destination devices.
Due to process variations inherent in the manufacturing process of ICs, individual ICs designed and intended to perform the same function can vary significantly. As a result, it is difficult to manufacture output drivers 14 with consistent output impedance. For example, the doping level, the length of channels in FETs, the thickness of the gate oxide for transistors, the diffusion resistance, and other characteristics associated with each individual IC vary during the manufacturing process. In other words, two supposedly identical ICs can vary in all of these characteristics. As these characteristics approach the ideal case the resistance of many components within a chip decrease. In the opposite extreme, as the IC characteristics stray further and further from the ideal case, the performance of the circuit degrades. Specifically, the resistance of the many components within the chip increases, which slows the response time of the IC.
In addition to manufacturing variation, a number of environmental factors may adversely affect IC performance. For example, supply voltage and ambient temperature can adversely effect individual ICs. More specifically, when an IC""s temperature approaches a maximum operating temperature the resistance of the FETs in the IC increases, which in turn leads to a decrease in efficiency and slower response times. Furthermore, when an IC""s supply voltage sags, as may occur in tandem with increases in ambient temperature, the IC""s response times may slow further.
One prior art approach to address operational problems introduced by manufacturing process variation and environmental factors such as supply voltage and ambient temperature is to provide a programmable output driver stage in a CMOS output driver. One such circuit is illustrated in FIG. 4A. The figure shows dual impedance controls 41 (e.g., programmable current sources, which serve to control the impedance of FETs 52 and 58) and a dual output driver 50. The dual output driver 50 drives a capacitively terminated transmission line 48. The transmission line 48 may be further identified by a characteristic impedance of xe2x80x9cZo.xe2x80x9d As illustrated in FIG. 4A, the pads 22, the capacitively terminated transmission line 48, the various electrical conductors connecting these an any other elements in a signal transmission path may be modeled as a composite transmission line 148.
Returning to the CMOS output driver, the dual impedance controls 41 determine a composite source impedance for the dual output driver 50. The composite source impedance can be separated into a value RSC (the source resistance while charging) and a value RSD (the source resistance while discharging). Generally speaking, it is desirable that RSC and RSD be equal to each other and to the characteristic impedance Zo of the transmission line 48, although one can imagine that there might be special circumstances that would require them to be different.
Note the capacitive load 49 at the destination end of transmission line 48. The system may employ a well-understood technique of doubling the output voltage by using reflected power from the reactive (and non-power dissipative) discontinuity (i.e., the capacitive load 49) at the terminus of the transmission line 48. It is desirable to achieve the full doubling effect without added overshoot (i.e., the case where Zo is too low, which can lead to multiple reflections) or excessive rise time (i.e., the case where Zo is too high, which can lead to multiple reflections). Note that when the load is reactive, the power that is launched by charging through RSC is transmitted through Zo, reflected (i.e., the load voltage is doubled), transmitted back through Zo and then absorbed by discharging, without re-reflection by the source resistance, RSC. A similar sequence of events occurs for discharging involving RSD. (All provided, of course, that RSC=Zo=RSD.) Yet even in a situation where there is a resistive termination with the expectation of genuine power transfer to the load without reflection, it is still important to control the source impedance of the output driver stages.
To appreciate the operation of the CMOS device, consider output driver stage 50, which includes four CMOS devices 52, 54, 56, and 58 connected as shown. Devices 54 and 56 act as switches to respectively pull-up (charge to VDD) and pull-down (discharge to GND) the signal that drives the transmission line 48 whose characteristic impedance, Zo, is to be matched by RSC (during pull-up) and by RSD (during pull-down). It will be understood that switching devices 54 and 56 are driven xe2x80x9conxe2x80x9d and xe2x80x9coffxe2x80x9d in a suitable scheme in accordance with the desired output waveform, and that although both devices 54 and 56 may be xe2x80x9coffxe2x80x9d simultaneously, both devices will never be on at the same time. In this regard, driver circuits 42 and 43 are provided to turn the switching devices 54 and 56 xe2x80x9conxe2x80x9d and xe2x80x9coff.xe2x80x9d Generally, and as is known, one driver circuit 42 operates to control the FET 54 to drive the output signal from a low to high value, while a second driver circuit 43 operates to control the FET 56 to drive the output signal from a high to low value.
Device 52 acts as a resistance of programmable value to combine with the very low on resistance of device 54 to produce RSC. Similarly, device 58 acts as a resistance of programmable value to combine with the relatively low xe2x80x9conxe2x80x9d resistance of device 56 to produce RSD. The resistance of device 52 is controlled by the value of the voltage PGATE 53, while in similar fashion the resistance of device 58 is determined by the value of the voltage NGATE 59. Assuming now that the P-type device 52 and N-type device 58 have generally equal transconductance, the signals NGATE 59 and PGATE 53 are controlled such that they (1) can be externally varied to adjust RSC and RSD over a suitably wide range of Zo despite process variations; (2) vary together such that as NGATE increases from GND toward VDD, PGATE 53 decreases correspondingly from VDD toward GND; and (3) automatically adjust to compensate for the effects of temperature.
Another common approach used to address operational problems introduced by manufacturing process variation and environmental factors is to configure multiple fingers in parallel using a digital logic scheme. One such circuit is illustrated in FIG. 4B. The figure shows dual digital impedance controls 41xe2x80x2, which serve to control the on/off state of NAND gates 61 or NOR gates 65. The on/off state of the NAND gates and NOR gates 65, together with the logic level of the input data, form an output driver suited to drive a capacitively terminated transmission line 48. The transmission line 48 may be further identified by a characteristic impedance of xe2x80x9cZo.xe2x80x9d As illustrated and explained with regard to the analog approach introduced in the circuit of FIG. 4A, the pads 22, the capacitively terminated transmission line 48, the various electrical conductors connecting these an any other elements in a signal transmission path may be modeled in the digital approach as well as a composite transmission line 148.
Returning attention to the digital output driver, the dual impedance controls 41xe2x80x2 determine a composite source impedance for the output driver. As in the analog or continuous output drive approach of the circuit in FIG. 4A, the composite source impedance can be separated into a value RSC (the source resistance while charging) and a value RSD (the source resistance while discharging).
The output driver of FIG. 4B includes three pairs of matched devices 62, 64 connected as shown. Devices 62, 64 act as switches to respectively pull-up (charge to VDD) and pull-down (discharge to GND) the signal that drives the transmission line 48 whose characteristic impedance, Zo, is to be matched during pull-up and during pull-down. It will be understood that switching devices 62, 64 are driven xe2x80x9conxe2x80x9d and xe2x80x9coffxe2x80x9d in a suitable scheme in accordance with the desired output waveform, and that although both devices 62 and 64 may be xe2x80x9coffxe2x80x9d simultaneously, both devices will never be xe2x80x9conxe2x80x9d at the same time.
Operationally, the circuit of FIG. 4B functions as follows. When the input data is logically high and the corresponding impedance control input is high, the associated NAND gate 61 will drive its corresponding PFET 62 xe2x80x9con.xe2x80x9d For those times when the input data signal is logically low and the corresponding impedance control input is low, the associated NOR gate 65 will drive its corresponding NFET xe2x80x9con.xe2x80x9d By controllably turning xe2x80x9conxe2x80x9d and xe2x80x9coffxe2x80x9d the NAND gates 61 and the NOR gates 65, the output impedance of the output driver may be adjusted. It should be appreciated that the responsiveness of the digital implementation illustrated in FIG. 4B will be greater than that of the analog implementation of FIG. 4A as there is a single FET between the supply voltage, VDD, and signal ground vs. the analog case where 2 FETs appear between the supply voltage and signal ground. It should be further appreciated that the sensitivity of each of the discrete impedance levels that result from adding each additional FET is dependent on the length of each respective FET junction.
In order to meet the high-speed performance requirements of modern ICs and the systems they support it is desirable to produce ICs that can support fast data transition times. Unfortunately, as IC clock and data signal rates approach and transition through the ultra-high frequency (UHF) range of the radio frequency spectrum (i.e., from 300 MHz to 3 GHz) the transmission lines themselves may behave as circuit elements. More specifically, at these frequencies sections of transmission lines can be designed to provide an inductive or capacitive impedance in order to match a particular expected load to enable maximum power transfer. More importantly, as IC clock and data signal rates increase the transmission lines increasingly attenuate the clock and data signals.
One prior-art approach to overcoming the high-frequency attenuation inherent within printed circuit board traces 25 is to increase the strength of the output driver. While this approach results in faster transition times or edge rates, the approach is undesirable in that as driver strength is increased, the output driver output impedance strays significantly from the generally desired 50 Ohms (i.e., the characteristic impedance of a typical printed circuit board trace 25). The increase in the impedance mismatch leads to an increase in the magnitude of the reflected signals, which in turn increases the noise and ringing of the transmitted signal 30 (FIGS. 2 and 3).
Accordingly, there is a need for an improved circuit and method for addressing the inherent transmission line induced high-frequency attenuation while retaining the benefits of an output impedance matched output driver.
In light of the foregoing, the invention relates to a circuit and method that maintains the impedance matching characteristics of a common output driver while compensating for the high-frequency signal attenuation inherent in printed circuit board traces, line bonding conductors, and integrated circuit (IC) packages. In a preferred embodiment, the circuit includes a pre-emphasis driver configured in parallel with a standard output driver. The pre-emphasis driver is a low-impedance driver configured as a tri-statable device which mirrors a received logic input when in an xe2x80x9conxe2x80x9d state (i.e., the pre-emphasis driver output mirrors the same logical sense as the standard driver.) When the pre-emphasis driver is in an xe2x80x9coffxe2x80x9d state, no output signal is present from the pre-emphasis driver and the pre-emphasis driver provides a high-impedance to signals reflected from destination devices and/or the transmission line. In accordance with a preferred embodiment, the pre-emphasis driver is controlled by a pre-emphasis control signal configured such that the pre-emphasis driver can inject high-frequency components into a transmission line for a portion of a clock cycle. The pre-emphasis control signal is configured such that the pre-emphasis driver turns xe2x80x9conxe2x80x9d in close approximation with data signal transitions from the standard driver and is turned xe2x80x9coffxe2x80x9d before a reflected signal caused by the impedance mismatch between the pre-emphasis driver and downstream elements (i.e., the transmission line and the receiver) appears at the parallel driver output.
A method for providing high-frequency compensation for data and clock signals is also disclosed. In its broadest terms, the method can be described as: providing a data signal at the input of a matched-impedance driver; configuring a low-impedance driver in parallel with the matched-impedance driver; providing a control signal to the low-impedance driver such that the low-impedance driver is turned xe2x80x9conxe2x80x9d when the data signal transitions and turned xe2x80x9coffxe2x80x9d before an impedance mismatch induced reflected signal returns at the output of the matched-impedance driver. The method for providing high-frequency compensation for data and clock signals outlined above results in a faster transition at receiving devices, while maintaining an impedance match at the output of improved driver.
Other objects, features, and advantages of the present invention will become apparent to one skilled in the art upon examination of the following drawings and detailed description. It is intended that all such additional objects, features, and advantages be included herein within the scope of the present invention, as defined by the claims.